The present invention relates to an electric motor controller for controlling a synchronous motor without using any position sensors.
A brushless motor is a representative example of a synchronous motor having no position sensor. A method of driving such electric motor is known as a conventional one wherein an induced voltage is detected to detect the position of the rotor of a brushless motor and the brushless motor is driven on the basis of the induced voltage. One of brushless motor driving methods is a rectangular wave driving method wherein a rectangular wave current flows through a brushless motor, and another is a sine wave driving method wherein a sine wave current flows therethrough. In the rectangular wave driving method, the waveform of the current is rectangular. Hence, the rectangular wave driving method is inferior to the sine wave driving method in all of motor vibration and noise. In the sine wave driving method, the zero-cross point of the current of a motor is detected. The applied voltage or command frequency of the motor is feedback-controlled so that the phase difference between the motor current and the applied voltage obtained on the basis of the zero-cross point becomes a desired command value.
The above-mentioned sine wave driving method for an electric motor controller (hereinafter is referred to as motor controller) in accordance with a first prior art will be described below referring to FIG. 26 and FIG. 27. FIG. 26 is a block diagram showing the motor controller in accordance with the first prior art for the sine wave driving method. In FIG. 26, a DC voltage of a DC power source 101 is converted into an AC voltage by an inverter circuit 102 and supplied to a motor 103 via a motor current detection section 104. The current of the motor 3 is detected by the motor current detection section 104 and input to an inverter control section 105. In the inverter control section 105, a frequency set by a frequency setting section 106 is supplied to a wave generation section 107. The wave generation section 107 generates the rotation phase and wave form of a voltage to be applied to the motor 103. The detection output of the motor current detection section 104 is applied to a current zero-cross detection section 108. The current zero-cross detection section 108 detects the zero-cross point of the motor current.
FIG. 27A is a graph showing a rotation phase θ. FIG. 27B is a graph showing the relationship between a motor current Is and an applied voltage Vs in a cycle T which is the inverse of an output frequency fs. As shown in FIG. 27A, the output frequency fs of the frequency setting section 106 is converted into a time T (=1/fs), i.e., a cycle, by the wave generation section 107, thereby generating the rotation phase θ. Furthermore, the wave generation section 107 generates a reference sine wave on the basis of the rotation phase θ. An output command computing section 112 generates the command value of the applied voltage Vs on the basis of the wave of the generated reference sine wave and the amplitude of a voltage computed by an error voltage computing section 111. The command value is applied to the inverter circuit 102. Hence, the motor current Is flows as shown in the waveform graph of FIG. 27B, thereby generating a phase difference ø between the motor current Is and the applied voltage Vs. The motor current Is detected by the motor current detection section 104 is applied to the current zero-cross detection section 108. The current zero-cross detection section 108 detects the phase at the zero-cross point of the motor current Is. The phase at the zero-cross point is applied to a phase difference computing section 109. The phase difference computing section 109 detects the phase difference ø between the applied voltage Vs and the motor current Is. An adder 113 obtains an error between the output of a phase difference command section 110 and the output of the phase difference computing section 109. The error is amplified by an error voltage computing section 111, thereby obtaining the amplitude of the motor applied voltage Vs. The applied voltage Vs, i.e., the output of the output command computing section 112, is subjected to pulse-width modulation (PWM) and applied to switching devices of the inverter circuit 102, thereby driving the inverter circuit 102.
In the above-mentioned first prior art, the zero-cross point of the motor current is detected, and feedback control is carried out so that the phase difference between the motor current Is and the applied voltage Vs becomes a desired command value. The zero-cross point of the motor current Is can be detected once every electrical angle of 180 degrees per phase as shown in FIG. 27B. In the case of three phase currents, the zero-cross point can be detected once every electrical angle of 60 degrees. However, a detection delay owing to sample holding is large in the feedback control wherein the zero-cross point is detected every electrical angle of 60 degrees. This detection delay makes the operation of the motor unstable at a low speed in particular, thereby causing the loss of synchronization, and being apt to produce the problem of stopping the rotation of the motor.
A sine wave driving method in accordance with a second prior art is disclosed in the Transactions of the Institute of Electrical Engineers of Japan, Volume D117, No. 1, 1997 and in Japanese Laid-open Patient Application No. Hei 11-18483. In this driving method, a motor voltage equation represented by the winding resistances of a motor and inductances on d-q axes is prepared in advance. Then, the phase and rotation speed of the motor are estimated from the applied voltage and the actual voltage of the motor, and feedback control is carried out. This second prior art will be described below referring to FIG. 28.
FIG. 28 is a block diagram showing a motor controller in accordance with the second prior art for the sine wave driving method. In FIG. 28, the DC power source 101, the inverter circuit 102, the motor current detection section 104 and the motor 103 are similar to those of the first prior art. The operation of an inverter control section 114 will be described below. The switching devices of the inverter circuit 102 are controlled on the basis of a PWM command value generated by an output command computing section 115, thereby driving the motor 103. The current flowing through the motor 103 at this time is detected by the motor current detection section 104, thereby outputting a detected signal. From the detected signal of the motor current and an estimated phase θ, the motor current is converted on γ-δ coordinate axes by a γδ conversion section 116, thereby outputting currents Iy and Iδ. The coordinate axes are d-q axes estimated on a motor model 117. The motor model 117 solves a motor voltage equation on the basis of the converted current and voltage command values, thereby outputting the estimated values of the phase θ and rotation speed ω. A frequency setting section 118 outputs the rotation frequency command value of the motor. An adder 119 computes the error between the command value of the rotation frequency and the estimated value of the rotation speed ω generated by the motor model 117, and outputs the error to a current command section 120. The current command section 120 PI-controls the error of the adder 119, thereby generating a current command value on the γ-δ axes. An adder 121 computes the error between the current command value from the current command section 120 and the currents Iy and Iδ from the γδ conversion section 116, and outputs the error to a voltage command section 122. The voltage command section 122 PI-controls the error of the adder 121, thereby generating an applied voltage command value. This applied voltage command value is used for the output command computing section 115 again and three-phase-converted, thereby forming a PWM command value. This PWM command value is used to control the inverter circuit 102 again in the next control cycle.
In the second prior art wherein the d-q axes are estimated by using the motor model, feedback is carried out at every control cycle (for example, at every carrier cycle). Hence, this technology has the advantage that hunting owing to a detection delay hardly occurs. On the other hand, motor parameters, especially inductances, fluctuate significantly owing to the influence of temperature and load. For this reason, if an error occurs between an actual motor parameter and a model used in a controller, the result of the estimation of a phase or a rotation speed becomes different from an actual value. Consequently, the rotation of the motor becomes uncontrollable in the end, thereby resulting in loss of synchronization. Hence, in order to prevent the loss of synchronization, it is necessary to correct the parameter by changing the rotation speed, load or temperature. Furthermore, in order to control motors different in parameters, it is necessary to adjust the parameters. It is thus difficult to readily apply this control to motors different in parameters. Still further, this control requires a large amount of computations since a current minor loop is used. It is thus necessary to use an expensive microcomputer or DSP.
A sine wave driving method in accordance with a third prior art is disclosed in Japanese Laid-open Patient Application No. 2000-262089. In this driving method, reactive power supplied to a motor is detected, and feedback is carried out so that the value of the reactive power becomes a target value. This third prior art will be described below referring to FIG. 29.
FIG. 29 is a block diagram showing a motor controller in accordance with the third prior art for the sine wave driving method. In FIG. 29, the DC power source 101, the inverter circuit 102, the motor current detection section 104 and the motor 103 are similar to those of the first prior art. The operation of an inverter control section 123 will be described below. An output command computing section 124 generates a PWM command signal from a command value of the voltage to be applied to the motor (hereinafter is referred to as “motor applied voltage command value”) and feeds the PWM command signal to the inverter circuit 102, thereby controlling the switching devices of the inverter circuit 102 and driving the motor 103. At this time, the motor current detection section 104 detects a current flowing through the motor 103, and outputs a detected signal. A coordinate conversion section 125 decomposes the motor current into an active current and a reactive current on the basis of the detected signal. A frequency setting section 126 outputs the rotation frequency command value of the motor 103. A reactive power command section 127 outputs a reactive power command value from the rotation frequency command value, the motor applied voltage command value, the active current and the reactive current. A reactive power computing section 128 computes a reactive power detected value from the motor applied voltage command value and the reactive current detected value. An adder 129 computes the error between the reactive power command value and the reactive power detected value, and an error voltage computing section 130 generates an applied voltage compensation value on the basis of this error. A V/f conversion section 131 generates a motor reference voltage from the rotation frequency command value. An adder 132 adds the motor reference voltage and the applied voltage compensation value, thereby generating a motor applied voltage command value. The motor applied voltage command value is input to the output command computing section 124 again, thereby generating a PWM command value. This PWM command value is used to control the inverter circuit 102 again at the next control cycle.
In the third prior art wherein feedback control is carried out so that reactive power is commanded so as to become a predetermined value, the feedback control is carried out at every control cycle. Hence, the third prior art has the advantage that hunting owing to a detection delay hardly occurs, just as in the case of the second prior art. Since no current minor loop is used, the third prior art has the advantage that an amount of computations is small, unlike the case of the second prior art. However, since the voltage applied to the motor is almost proportional to the rotation speed of the motor, it is necessary to change the reactive power command value depending on the change in the rotation speed. Still further, since a motor parameter is used for the generation of the command value, correction is necessary depending on the change in the parameter, just as in the case of the second prior art. Hence, the computation of the reactive power command value becomes complicated. After all, the amount of all the computations is large, whereby it is necessary to use an expensive microcomputer or DSP. Just as in the case of the second prior art, in order to control motors different in parameters, it is necessary to adjust the parameters. It is thus difficult to readily apply this control to motors different in parameters. This prior art relates to a controlling wherein the output torque of the motor becomes its maximum value at all times, and in such controlling, the field-weakening control for use in the case of insufficient voltage cannot be carried out. Therefore, there is a problem that the range of the rotation speed is limited.